Receiver of a pulsed light signal with wide dynamic range

ABSTRACT

A receiver of a pulsed light signal comprises a photodiode adapted to generate an electric current in response to this light signal, having a parasitic capacitance C d  as its characteristic; an electrical ground; and a transimpedance amplifier connected to the input of the photodiode by a linking capacitor C liaison . It includes an attenuation pad located between the photodiode and the transimpedance amplifier, consisting of a capacitor C p  where C p =C d /(α−1), α being a predetermined attenuation, where α&gt;1.

The field of the invention is that of a photodiode receiver which receives light pulses over a very wide dynamic range (from several nanoamperes to several tens of milliamperes). This dynamic range is provided by means of a device for gain switching by discrete values, called “pads”.

The most commonly used solution for producing such a receiver is that of providing a photodiode with a TIA (an acronym for the English expression “Transimpedance Amplifier”); the sensitivity performance is dependent on this TIA, which has a high gain bandwidth product (or “GBW”, an acronym for the English expression “Gain Bandwidth Product”) and very low noise.

A photodiode 1 is conventionally represented by the circuit shown in FIG. 1 a. As shown in the left-hand figure, this photodiode is preferably charged by a resistor R′_(d) between the anode and the ground so as to absorb the direct current due to the ambient illumination, also called the background current, which is included in the light signal received by the photodiode. According to the equivalent representation shown in the right-hand figure, this resistor R′_(d) in parallel with the internal resistance of the photodiode forms an equivalent resistor Rd. The photodiode is generally characterized by a capacitance Cd between the anode and the ground, shown in the right-hand figure.

In a conventional receiver circuit, an example of which is shown in FIG. 1 b, a photodiode 1 of this type is associated with a TIA 2 via a linking capacitor C_(liaison) which helps to separate the useful pulses from the background current. The value of this linking capacitor C_(liaison) is typically more than 10 nF. Let us recall that a TIA comprises, in parallel, an operational amplifier AOP or an amplifier with discrete components, a feedback resistor R_(f) and a stabilizing capacitor C_(f). Such a receiver makes it possible to neutralize the effect of the parasitic capacitance C_(d) of the photodiode by means of a virtual ground.

To a first approximation, this is a second-order loop system:

-   -   Having a conversion gain Z_(T)(p) such that

$\begin{matrix} {{Z_{T}(p)} = {\frac{V_{s}}{i_{D}} = {{- R_{f}}\frac{1}{1 + {\frac{2\; \zeta}{\omega_{n}}p} + \frac{p^{2}}{\omega_{n}^{2}}}}}} & \left( {{eq}\mspace{14mu} 1} \right) \end{matrix}$

-   -   where V_(s) is the output voltage of the circuit, i_(D) is the         current generated by the photodiode, p (p=jω=j2πf) is the         Laplace variable, R_(f) is the feedback resistance of the TIA,         and ξ is the damping of the receiver,     -   and having a natural frequency ω_(n) such that:

$\begin{matrix} {\omega_{n} = \sqrt{\frac{2\; \pi \; {GBW}}{R_{f}\left( {C_{d} + C_{f}} \right)}}} & \left( {{eq}\mspace{14mu} 2} \right) \end{matrix}$

The ratio of damping to natural frequency can be written thus:

$\begin{matrix} {\frac{\zeta}{\omega_{n}} = {\frac{1}{2}\left( {{R_{f}C_{f}} + {\frac{1}{2\; \pi \; {GBW}}\left( {1 + \frac{R_{f}}{R_{d}}} \right)}} \right)}} & \left( {{eq}\mspace{14mu} 3} \right) \end{matrix}$

or in practice:

$\begin{matrix} {{R_{f}C_{f}}\operatorname{>>}{\frac{1}{2\pi \mspace{11mu} G\; B\; W}\left( {1 + \frac{R_{f}}{R_{d}}} \right)}} & \left( {{eq}\mspace{14mu} 4} \right) \end{matrix}$

this ratio then takes the simple form:

$\begin{matrix} {\frac{\zeta}{\omega_{n}} \approx {\frac{1}{2}R_{f}C_{f}}} & \left( {{eq}\mspace{14mu} 5} \right) \end{matrix}$

The gain modification is found according to Equation (1) from the change in the value R_(f) which, according to Equation (2), modifies the natural frequency ω_(n) and hence the damping ξ according to Equation (5). With a conventional solution, therefore, it appears to be difficult to change the gain without modifying the transfer function.

The frequency response is shown in FIG. 5 a for three damping values ξ (0.9, 0.7 and 0.5). This figure demonstrates that the change in gain affects the damping when the band is kept constant.

Another important criterion is the equivalent current noise applied to the input of the TIA, which is written thus:

$\begin{matrix} {i_{n} = \sqrt{i_{n -}^{2} + \left( \frac{e_{n}}{R_{f}} \right)^{2} + \frac{4{kT}}{R_{f}}}} & \left( {{eq}\mspace{14mu} 6} \right) \end{matrix}$

where i_(n−) and e_(n), respectively, are the equivalent noise current at the negative input of the operational amplifier AOP and the equivalent noise voltage at the input of AOP which characterize the operational amplifier used, k is the Boltzmann constant, and T is the temperature in degrees Kelvin.

For a given TIA and a given photodiode, the sensitivity is optimized by choosing the highest possible resistance R_(f) compatible with the pulse processing band.

However, as the gain increases, the admittance decreases, because the voltage range at the output of the amplifier is fixed by the power supplies. Conversely, a decrease in gain increases the admittance but degrades the noise, with a current limitation determined by the maximum output current of the amplifier.

The problem therefore arises of providing an optimum receiver for weak signals but also for strong signals, while preferably maintaining the same frequency response. The conventional solutions are:

-   -   Reducing the gain of the TIA by reducing the feedback resistor         R_(f) which determines the conversion gain of the TIA, thereby         improving the admittance but worsening the noise. Furthermore,         reducing the feedback resistance has the effect of significantly         increasing the bandwidth, which is evidently undesirable if a         pulse shape independent of gain is required.     -   Placing a switched resistive attenuator between the photodiode         and the TIA so as to reduce the gain when the received level         exceeds the admittance. This degrades the noise, because the         resistances generate noise. Moreover, the switches have         non-negligible parasitic capacitance relative to the capacitance         of the photodiode, which affects the transfer function.

The conventional solutions do not meet the requirement.

Consequently there is still a need for a receiver with a wide dynamic range, optimized in terms of noise.

More precisely, the invention proposes a receiver of a pulsed light signal comprising:

-   -   a photodiode adapted to generate an electric current I_(d) in         response to the light signal, having a parasitic capacitance         C_(d) as its characteristic,     -   an electrical ground, and     -   a transimpedance amplifier connected to the input of the         photodiode by a linking capacitor C_(liaison).

It is primarily characterized in that it includes a series-parallel reactive circuit, consisting of a capacitor C_(p) which, combined with the diode capacitance C_(d), forms a current divider, called an attenuation pad, upstream of the transimpedance amplifier.

This current divider enables the signal to be attenuated without degrading the noise.

The capacitor Cp is typically placed in series with the linking capacitor and generally supplements it.

According to one characteristic of the invention, the receiver includes a background current resistor R_(d) located between the photodiode and the electrical ground, the capacitance C_(d) and said resistor R_(d) having an impedance Z_(d), and the attenuation pad also consists of a resistor R_(p) in parallel with the capacitor C_(p), thus forming a parallel electrical network called an aperiodic attenuation pad, having an impedance Z_(p), where

Z _(p)=(α−1)Z _(d.)

This aperiodic attenuation pad can be used to compensate the effect of the resistor Rd and thereby maintain the low frequency response of the receiver.

If required, the attenuation pad further comprises a switch in parallel with the capacitor C_(p), so as to produce a switchable attenuation pad. This switch enables the circuit R_(p) C_(p) to be short-circuited or switched.

The attenuation pad may also include a capacitor C_(opt) in parallel with C_(d), this capacitor C_(opt) itself being switchable if required.

The aperiodic attenuation pad may also comprise a compensation capacitor C_(comp) in parallel with the input of the transimpedance amplifier, thus forming a compensated aperiodic attenuation pad, with C_(comp)=C_(d)(α−1)/α, this compensation capacitor being switchable if required.

Given that the assembly consisting of the attenuation pad and the transimpedance amplifier is called a receiving channel with attenuation pad, the receiver further comprises a receiving channel without attenuation pad, comprising another transimpedance amplifier, these receiving channels being multiplexed by means of an input switch of these channels and an output switch of these channels, the switches being synchronized with one another so as to produce a receiver with different gains. Evidently, other receiving channels with attenuation pads may be multiplexed with said receiving channels, each receiving channel with an attenuation pad having a different attenuation.

The light signal is typically capable of generating current pulses in the range from 10 nA to 100 mA in the photodiode.

Other characteristics and advantages of the invention will be revealed by the following detailed description, provided by way of non-limiting example, with reference to the attached drawings, in which:

FIG. 1 a, described above, shows two equivalent schematic representations of a photodiode having a background resistor;

FIG. 1 b shows schematically a receiver circuit according to the prior art, including a photodiode and a TIA;

FIG. 2 shows schematically an example of a receiver circuit according to a first embodiment of the invention with a purely capacitive attenuation pad;

FIG. 3 a shows schematically an example of a receiver circuit according to a second embodiment of the invention, with an aperiodic attenuation pad switched around Rp Cp;

FIG. 3 b shows schematically an example of a receiver circuit according to a third embodiment of the invention, with an aperiodic attenuation pad switched around Cd;

FIG. 3 c shows schematically an example of a receiver circuit according to a fourth embodiment of the invention, with a compensated aperiodic attenuation pad;

FIG. 4 shows schematically an example of a receiver circuit according to a fifth embodiment of the invention, with a plurality of switched receiving channels;

FIG. 5 a shows the frequency response of a conventional receiver with constant bandwidth for three values of gain, showing a variation in damping;

FIG. 5 b shows the frequency response of a receiver with a compensated aperiodic pad with a constant bandwidth for three values of gain, showing how the damping is maintained.

The same elements are identified by the same references in all the figures.

The receiver according to the invention is based on the principle of a current divider bridge which is capacitive instead of resistive.

An example of a capacitive attenuation pad associated with a photodiode 1 equipped with a TIA 2 is shown in FIG. 2. In this figure, the aim is more particularly to indicate the electrical currents.

The photodiode is an ideal current generator, and is capacitive because of the parasitic capacitance Cd. When a capacitor Cp is added in series between the TIA 2 and the photodiode 1, at the input or output of the linking capacitor, the current generated by the photodiode is distributed between the capacitance Cd and the capacitor Cp as a function of the values of the capacitances:

$I_{F} = \frac{I_{D}}{\alpha}$

via the capacitance C_(P);

$I_{Cd} = {\frac{\alpha - 1}{\alpha}I_{D}}$

via the capacitance C_(d); The value of the capacitance

$C_{p} = \frac{C_{d}}{\alpha - 1}$

determines the attenuation

$\alpha = \frac{C_{d} + C_{p}}{C_{p}}$

of the capacitive divider. The signal is therefore attenuated without the addition of supplementary noise.

We find that α>1; in practice, an attenuation α typically in the range from 2 to 30 is chosen. The value of C_(p) is typically less than 10 pF.

This attenuation pad 30 consisting of the capacitor Cp is provided, if required, with a switch 31 placed in parallel with this capacitor Cp to adapt the gain to the received level.

Let us analyze in greater detail the behavior of such a receiver at low frequencies, that is to say below 100 kHz:

As indicated in the preamble, the photodiode 1 is generally charged by a resistor Rd so as to absorb the direct current due to the ambient illumination. This resistor Rd modifies the impedance of the photodiode, which can then no longer be considered as purely capacitive.

As shown in FIG. 3 a, the capacitor Cp is then supplemented with a resistor Rp in parallel, which forms, with this capacitor, a parallel electrical network called an aperiodic attenuation pad 30 having an impedance Zp, proportional to Zd which is the impedance of the diode circuit including the resistance R_(d) and the capacitor C_(d) in parallel.

Assuming that Zp=(α−1)Zd, we find:

$\quad\left\{ \begin{matrix} {\; {I_{F} = \frac{I_{D}}{\alpha}}} \\ {R_{p} = {\left( {\alpha - 1} \right)R_{d}}} \\ {C_{p} = {\frac{1}{\left( {\alpha - 1} \right)}C_{d}}} \end{matrix} \right.$

I_(F) being the output current of the attenuation pad 30.

The attenuation of the current then becomes independent of frequency, the additional noise remaining very low because the resistor Rp is large relative to Rd, owing to the attenuation ratio α.

This aperiodic attenuation pad 30 is provided, if required, with a switch 31 placed in parallel with Rp and Cp.

Let us now analyze in greater detail the behavior of such a receiver at high frequencies, that is to say above 10 MHz:

With the previous receiver circuit, the TIA 2 no longer sees the same impedance when the attenuation pad is active, and its transfer function is affected by this, as shown in FIG. 5 a for curves of gain as a function of frequency for three values of damping ξ(0.9, 0.7 and 0.5). The circuit behaves as a second-order system.

The ratio of damping to natural frequency is:

$\frac{\hat{\zeta}}{{\hat{\omega}}_{n}} = {\frac{1}{2} \cdot \left\lbrack {{R_{f} \cdot C_{f}} + {\frac{1}{{2 \cdot \pi \cdot G}\; B\; W} \cdot \left( {1 + \frac{R_{f}}{a \cdot R_{d}}} \right)}} \right\rbrack}$

When the condition of a sufficient product of gain×band is met:

${R_{f} \cdot C_{f}}\operatorname{>>}{\frac{1}{2\pi \mspace{11mu} G\; B\; W}\left( {1 + \frac{R_{f}}{a\mspace{11mu} R_{d}}} \right)}$

the ratio of damping to natural frequency remains constant:

$\frac{\hat{\zeta}}{{\hat{\omega}}_{n}} = {\frac{\zeta}{\omega_{n}} \cong {\frac{1}{2}R_{f}C_{f}}}$

But:

-   -   The natural frequency {circumflex over (ω)}_(n) corresponds to         that of a circuit whose photodiode has a parasitic capacitance         which is reduced by a ratio α:

$\omega_{n} = {\left. \sqrt{\frac{2\pi \mspace{11mu} G\; B\; W}{R_{f}\left( {C_{d} + C_{f}} \right)}}\Rightarrow{\hat{\omega}}_{n} \right. = \sqrt{\frac{2\pi \mspace{11mu} G\; B\; W}{R_{f}\left( {\frac{C_{d}}{\alpha} + C_{f}} \right)}}}$

-   -   The static gain Z_(T) is divided by α, as desired:

$Z_{T} = {\left. {{- R_{f}}\frac{1}{1 + {\frac{2\zeta}{\omega_{n}}p} + \frac{p^{2}}{\omega_{n}^{2}}}}\Rightarrow{\hat{Z}}_{T} \right. = {\frac{- R_{f}}{\alpha} \times \frac{1}{1 + {\frac{2\hat{\zeta}}{{\hat{\omega}}_{n}}p} + \frac{p^{2}}{{\hat{\omega}}_{n}^{2}}}}}$

Since an attenuation α is created, the natural frequency {circumflex over (ω)}_(n) of the receiver also increases, but the damping increases because the ratio of damping to natural frequency remains constant.

To retain the same bandwidth with and without attenuation, the damping must be modified; compensation is therefore added to produce the same transfer function.

Since the ratio of damping to natural frequency is invariant, the damping and the natural frequency are maintained simultaneously by adding a compensation capacitor C_(comp) 43 shown in FIG. 3 c, in parallel on the input of the TIA 2, such that:

$\omega_{n} = {\left. {\hat{\omega}}_{n}\Leftrightarrow\sqrt{\frac{2\pi \mspace{11mu} G\; B\; W}{R_{f}\left( {\frac{C_{d}}{\alpha} + C_{COMP} + C_{f}} \right)}} \right. = \sqrt{\frac{2\pi \mspace{11mu} G\; B\; W}{R_{f}\left( {C_{d} + C_{f}} \right)}}}$ Therefore: $C_{comp} = {\frac{\alpha - 1}{\alpha}C_{d}}$

The aperiodic attenuation pad modified in this way is then called a “compensated aperiodic attenuation pad”.

Such a receiver exhibits the same transfer function regardless of whether or not the pad is active.

In addition to the switch 31 (the first switch), another switch 44 may be placed in series with the compensation capacitor C_(comp), between the latter and the ground. The compensated aperiodic attenuation pad 30 operates when this other switch 44 is closed and the first switch 31 is open, and vice versa.

In the definition of the aperiodic pad, the value of the capacitor C_(p) is related to the capacitance C_(d) of the detector and to the attenuation ratio. For a value of Cd in the range from 12 to 18 pF, we therefore find, according to the formula

${Cp} = {\frac{1}{\left( {\alpha - 1} \right)}{Cd}}$

and with α in the range from 10 to 20, a very low value of Cp in the range from 0.5 to 2 pF, which is difficult to control in an industrial context in the production of a circuit. The solution proposed in FIG. 3 b consists in artificially increasing the capacitance Cd by adding a capacitor C_(opt) 41 in parallel, thereby enabling the value of Cp to be increased at an equal attenuation. This capacitor C_(opt) can be switched by a switch 42 placed in series toward the ground.

In practice, switches are imperfect, and fitting them may introduce parasitic elements which, in some cases, may degrade the transfer function. The term “receiving channel with an attenuation pad 50” denotes the assembly consisting of the attenuation pad 30 and the transimpedance amplifier 2. The attenuation pad may or may not be aperiodic, may or may not be switchable, may or may not be compensated, and so forth. A proposed alternative is to use a plurality of receiving channels, each having a different gain, as shown in FIG. 4 with two values of gain. In this example, the receiver has two receiving channels:

-   a receiving channel 50 with a pad, optimized with a compensated     aperiodic attenuation pad, and -   a receiving channel 50′ without a pad (having only a transimpedance     amplifier 2) optimized at maximum gain.

The channel is typically selected by means of a switch 61 located at the input of these channels and a switch 62 located at the output of these channels, these switches being synchronized with one another to produce a receiver with different gains. The input switch 61 is advantageously provided with a linking capacitor on each of its outputs leading to a receiving channel.

The receiver provided with an attenuation pad in this way has the following advantages:

-   -   Greater admittance than a conventional circuit;     -   A frequency response independent of the gain;     -   Optimized noise;     -   Allowance for the parasitic capacitances of the switches;     -   No need for a compromise between sensitivity and power behavior;     -   Simplicity of production.

This receiver is typically integrated into a Lidar system. It may be used as an element of a distance gauge, notably a semi-active distance gauge, that is to say one equipped with a designation laser adapted to illuminate a target whose backscatter is measured by this receiver. The target emits, for example, light pulses at a constant level, but if the receiver is at a long distance it can only measure very low-level pulses, whereas it can measure high-level pulses when it is at a short distance. 

1. A receiver of a pulsed light signal comprising: a photodiode adapted to generate an electric current (I_(D)) in response to this light signal, having a capacitance C_(d) as its characteristic, an electrical ground, and a transimpedance amplifier connected to the input of the photodiode by a linking capacitor C_(liaison), and an attenuation pad located between the photodiode and the transimpedance amplifier, consisting of a capacitor C_(p) where C_(p)=C_(d)/(α−1), a being a predetermined attenuation, where α>1.
 2. The receiver of a light signal as claimed in claim 1, wherein the capacitor C_(p) is placed in series with the linking capacitor C_(liaison).
 3. The receiver of a light signal as claimed in claim 1, wherein the linking capacitor is integrated with the capacitor C_(p).
 4. The receiver of a pulsed light signal as claimed in claim 1, comprising a background current resistor R_(d) located between the photodiode and the electrical ground, the capacitance C_(d) and said resistor R_(d) having an impedance Z_(d), and in that the attenuation pad also consists of a resistor R_(p) in parallel with the capacitor C_(p), thus forming a parallel electrical network called an aperiodic attenuation pad, having an impedance Z_(p), where Z_(p)=(α−1)Z_(d).
 5. The receiver of a pulsed light signal as claimed in claim 1, wherein the attenuation pad further includes a switch in parallel with the capacitor C_(p), so as to produce a switchable attenuation pad.
 6. The receiver of a pulsed light signal as claimed in claim 1, wherein the attenuation pad further includes a capacitor C_(opt) in parallel with C_(d).
 7. The receiver of a pulsed light signal as claimed in claim 6, wherein the attenuation pad further includes a switch in series with the capacitor C_(opt).
 8. The receiver of a pulsed light signal as claimed in claim 1, considered in combination with claim 4, wherein the aperiodic attenuation pad also consists of a compensation capacitor C_(comp) in parallel with the input of the transimpedance amplifier, thus forming a compensated aperiodic attenuation pad, with C_(comp)=C_(d)(α−1)/α.
 9. The receiver of a pulsed light signal as claimed in claim 8, considered in combination with claim 5, further comprising a switch in series with the compensation capacitor and connected to the ground, so as to produce a switchable compensation capacitor.
 10. The receiver of a pulsed light signal as claimed in claim 1, wherein, the assembly consisting of the attenuation pad and the transimpedance amplifier being called a receiving channel with an attenuation pad, the receiver further comprises a receiving channel without an attenuation pad, comprising another transimpedance amplifier, these receiving channels being multiplexed by means of an input switch of these channels and an output switch of these channels, the switches being synchronized with one another so as to produce a receiver with different gains.
 11. The receiver of a pulsed light signal as claimed in claim 10, comprising at least another receiving channel with an attenuation pad, multiplexed with said receiving channels, each receiving channel with an attenuation pad having a different attenuation.
 12. The receiver of a pulsed light signal as claimed in claim 1, wherein the transimpedance amplifier includes an operational amplifier or an amplifier with discrete components.
 13. The receiver of a pulsed light signal as claimed in claim 1, wherein the light signal is capable of generating current pulses in the range from 10 nA to 100 mA in the photodiode.
 14. The receiver of a pulsed light signal as claimed in claim 1, wherein α is in the range from 2 to
 30. 15. A Lidar including a receiver of a pulsed light signal as claimed in claim
 1. 16. A distance gauge equipped with a receiver as claimed in claim
 1. 17. The distance gauge as claimed in claim 16, further equipped with a designation laser. 